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functional block diagram ldac span/ bip off agnd db15 (msb) (lsb) db0 dgnd cs l1 ref in ref out ? ee +v cc +v ll v out 16-bit latch 10k amp AD669 22 7 6 5 23 27 28 10v ref 2 14 3 26 24 10k 10.05k 16-bit latch 16-bit dac 25 rev. a information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a monolithic 16-bit dacport AD669 features complete 16-bit d/a function on-chip output amplifier high stability buried zener reference monolithic bimos ii construction 6 1 lsb integral linearity error 15-bit monotonic over temperature microprocessor compatible 16-bit parallel input double-buffered latches fast 40 ns write pulse unipolar or bipolar output low glitch: 15 nv-s low thd+n: 0.009% mil-std-883 compliant versions available general description the AD669 dacport ? is a complete 16-bit monolithic d/a converter with an on-board reference and output amplifier. it is manufactured on analog devices bimos ii process. this pro- cess allows the fabrication of low power cmos logic functions on the same chip as high precision bipolar linear circuitry. the AD669 chip includes current switches, decoding logic, an output amplifier, a buried zener reference and double-buffered latches. the AD669s architecture insures 15-bit monotonicity over temperature. integral nonlinearity is maintained at 0.003%, while differential nonlinearity is 0.003% max. the on-chip output amplifier provides a voltage output settling time of 10 m s to within 1/2 lsb for a full-scale step. data is loaded into the AD669 in a parallel 16-bit format. the double-buffered latch structure eliminates data skew errors and provides for simultaneous updating of dacs in a multi-dac system. three ttl/lsttl/5 v cmos compatible signals con- trol the latches: cs , l1 and ldac. the output range of the AD669 is pin programmable and can be set to provide a unipolar output range of 0 v to +10 v or a bipolar output range of C10 v to +10 v. the AD669 is available in seven grades: an and bn versions are specified from C40 c to +85 c and are packaged in a 28-pin plastic dip. the ar and br versions are specified for C40 c to +85 c operation and are packaged in a 28-pin soic. the sq version is specified from C55 c to +125 c and is packaged in a hermetic 28-pin cerdip package. the AD669 is also available compliant to mil-std-883. refer to the AD669/883b data sheet for specifications and test conditions. dacport is a registered trademark of analog devices, inc. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 617/329-4700 fax: 617/326-8703 product highlights 1. the AD669 is a complete voltage output 16-bit dac with voltage reference and digital latches on a single ic chip. 2. the internal buried zener reference is laser trimmed to 10.000 volts with a 0.2% maximum error. the reference voltage is also available for external applications. 3. the AD669 is both dc and ac specified. dc specs include 1 lsb inl error and 1 lsb dnl error. ac specs include 0.009% thd+ n and 83 db snr. the ac specifications make the AD669 suitable for signal generation applications. 4. the double-buffered latches on the AD669 eliminate data skew errors while allowing simultaneous updating of dacs in multi-dac systems. 5. the output range is a pin-programmable unipolar 0 v to +10 v or bipolar C10 v to +10 v output. no external compo- nents are necessary to set the desired output range. 6. the AD669 is available in versions compliant with mil- std-883. refer to the analog devices military products databook or current AD669/883b data sheet for detailed specifications.
AD669Cspecifications model AD669an/ar AD669aq/sq AD669bn/bq/br min typ max min typ max min typ max units resolution 16 16 16 bits digital inputs (t min to t max ) v ih (logic 1 ) 2.0 5.5 * * * * volts v il (logic 0 ) 0 0.8 * * * * volts i ih (v ih = 5.5 v) 6 10 ** m a i il (v il = 0 v) 6 10 ** m a transfer function characte ristics 1 integral nonlinearity 6 2 * 6 1 lsb t min to t max 6 4 * 6 2 lsb differential nonlinearity 6 2 * 6 1 lsb t min to t max 6 4 * 6 2 lsb monotonicity over temperature 14 14 15 bits gain error 2, 5 6 0.15 6 0.10 6 0.10 % of fsr gain drift 2 (t min to t max ) 25 15 15 ppm/ c unipolar offset 6 5 6 5 6 2.5 mv unipolar offset drift (t min to t max ) 533 ppm/ c bipolar zero error 6 15 6 15 6 10 mv bipolar zero error drift (t min to t max ) 12 10 5 ppm/ c reference input input resistance 7 10 13 ****** k w bipolar offset input resistance 7 10 13 ****** k w reference output voltage 9.98 10.00 10.02 ****** volts drift 25 15 15 ppm/ c external current 3 24 ** ** ma capacitive load 1000 * * pf short circuit current 25 * * ma output characteristics output voltage range unipolar configuration 0 +10 * * * * volts bipolar configuration C10 +10 * * * * volts output current 5 * * ma capacitive load 1000 * * pf short circuit current 25 * * ma power supplies voltage v cc 4 +13.5 +16.5 * * * * volts v ee 4 C13.5 C16.5 * * * * volts v ll +4.5 +5.5 * * * * volts current (no load) i cc +12 +18 ** ** ma i ee C12 C18 ** ** ma i ll @ v ih , v il = 5, 0 v 0.3 2 ** ** ma @ v ih , v il = 2.4, 0.4 v 3 7.5 ** ** ma power supply sensitivity 1 3 * * * * ppm/% power dissipation (static, no load) 365 625 * * mw temperature range specified performance (a, b) C40 +85 C40 +85 C40 +85 c specified performance (s) C55 +125 c notes 1 for 16-bit resolution, 1 lsb = 0.0015% of fsr = 15 ppm of fsr. for 15-bit resolution, 1 lsb = 0.003% of fsr = 30 ppm of fsr. for 14-bit resolution 1 lsb = 0.006% of fsr = 60 ppm of fsr. fsr stands for full-scale range and is 10 v for a 0 v to + 10 v span and 20 v for a C10 v to +10 v span. 2 gain error and gain drift measured using the internal reference. gain drift is primarily reference related. see the using the AD669 with the ad688 reference section for further information. 3 external current is defined as the current available in addition to that supplied to ref in and span/bipolar offset on the AD669. 4 operation on 12 v supplies is possible using an external reference like the ad586 and r educing the output range. refer to the internal/external reference use section. 5 measured with fixed 50 w resistors. eliminating these resistors increases the gain error by 0.25% of fsr (unipolar mode) or 0.50% of fsr (bipolar mode). refer to the analog circuit connections section. *same as AD669an/ar specification. specifications subject to change without notice. specifications in boldface are tested on all production units at final electrical test. results from those tests are used to calculate outgoing quality levels. all min and max specifica- tions are guaranteed. those shown in boldface are tested on all production units. (@ t a = +25 8 c, v cc = +15 v, v ee = C15 v, v ll = +5 v, unless otherwise noted) rev. a C2C
AD669 rev. a C3C timing characteristics v cc = +15 v, v ee = C15 v, v ll = +5 v, v hi = 2.4 v, v lo = 0.4 v limit limit limit C40 8 c to C55 8 c to parameter +25 8 c +85 8 c +125 8 c units (figure la) t cs 40 50 55 ns min t li 40 50 55 ns min t ds 30 35 40 ns min t dh 10 10 15 ns min t lh 90 110 120 ns min t lw 40 45 45 ns min (figure lb) t low 130 150 165 ns min t high 40 45 45 ns min t ds 120 140 150 ns min t dh 10 10 15 ns min specifications subject to change without notice. specifications in boldface are tested on all production units at final electrical test. results from those tests are used to calculate outgoing quality levels. all min and max specifications are guaranteed. those shown in boldface are tested on all production units. ac performance characteristics parameter limit units test conditions/comments output settling time 13 m s max 20 v step, t a = +25 c (time to 0.0008% fs 8 m s typ 20 v step, t a = +25 c with 2 k w , 1000 pf load) 10 m s typ 20 v step, t min t a t max 6 m s typ 10 v step, t a = +25 c 8 m s typ 10 v step, t min t a t max 2.5 m s typ 1 lsb step, t min t a t max total harmonic distortion + noise a, b, s grade 0.009 % max 0 db, 1001 hz; sample rate = 100 khz; t a = +25 c a, b, s grade 0.07 % max C20 db, 1001 hz; sample rate = 100 khz; t a = +25 c a, b, s grade 7.0 % max C60 db, 1001 hz; sample rate = 100 khz; t a = +25 c signal-to-noise ratio 83 db min t a = +25 c digital-to-analog glitch impulse 15 nv-s typ dac alternately loaded with 8000h and 7fffh digital feedthrough 2 nv-s typ dac alternately loaded with 0000h and ffffh; cs high output noise voltage 120 nv/ ? hz typ measured at v out , 20 v span; excludes reference density (1 khz C 1 mhz) reference noise 125 nv/ ? hz typ measured at ref out specifications subject to change without notice. specifications in boldface are tested on all production units at final electrical test. results from those tests are used to calculate outgoing quality levels. all min and max specifications are guaranteed. those shown in boldface are tested on all production units. (with the exception of total harmonic distortion + noise and signal-to-noise ratio, these characteristics are included for design guidance only and are not subject to test. thd+n and snr are 100% tested. t min t a t max , v cc = +15 v, v ee = C15 v, v ll = +5 v except where noted.) data ldac t ds t dh cs t lw t lh l1 t cs t l1 figure 1a. AD669 level triggered timing diagram data t ds t dh cs and/or l1 , ldac tie cs and/or l1 to ground or together with ldac t low t high figure 1b. AD669 edge triggered timing diagram
AD669 rev. a C4C ordering guide linearity gain temperature error max tc max package package model range t min Ct max ppm/ 8 c description option* AD669an C40 c to +85 c 4 lsb 25 plastic dip n-28 AD669ar C40 c to +85 c 4 lsb 25 soic r-28 AD669bn C40 c to +85 c 2 lsb 15 plastic dip n-28 AD669br C40 c to +85 c 2 lsb 15 soic r-28 AD669aq C40 c to +85 c 4 lsb 15 cerdip q-28 AD669bq C40 c to +85 c 2 lsb 15 cerdip q-28 AD669sq C55 c to +125 c 4 lsb 15 cerdip q-28 AD669/883b** C55 c to +125 c** ** ** ** * * n = plastic dip; q = cerdip; r = soic. ** refer to AD669/883b military data sheet. esd sensitivity the AD669 features input protection circuitry consisting of large transistors and polysilicon series resistors to dissipate both high-energy discharges (human body model) and fast, low-energy pulses (charged device model). per method 3015.2 of mil-std-883: c, the AD669 has been classified as a class 2 device. proper esd precautions are strongly recommended to avoid functional damage or performance degradation. charges as high as 4000 volts readily accumulate on the human body and test equipment and discharge without detection. unused devices must be stored in conductive foam or shunts, and the foam should be discharged to the destination socket before devices are removed. for further information on esd precautions, refer to analog devices esd prevention manual. warning! esd sensitive device absolute maximum ratings * v cc to agnd . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +17.0 v v ee to agnd . . . . . . . . . . . . . . . . . . . . . . . +0.3 v to C17.0 v v ll to dgnd . . . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v agnd to dgnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 v digital inputs (pins 5 through 23) to dgnd . . . . . . C1.0 v to . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7.0 v ref in to agnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10.5 v span/bipolar offset to agnd . . . . . . . . . . . . . . . . . . . 10.5 v ref out, v out . . . . . . indefinite short to agnd, dgnd, . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . v cc , v ee , and v ll power dissipation (any package) to +60 c . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1000 mw derates above +60 c . . . . . . . . . . . . . . . . . . . . . .8.7 mw/ c storage temperature . . . . . . . . . . . . . . . . . . . C65 c to +150 c lead temperature (soldering, 10 sec) . . . . . . . . . . . . . +300 c *stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only, and functional operation of the device at these or any other conditions above those indi cated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. pin configuration db15 db14 db13 db12 db11 db10 db9 db8 dgnd v ee v cc v ll cs l1 1 2 3 4 5 6 7 8 9 10 11 12 13 14 28 27 26 25 24 23 22 21 20 19 18 17 16 15 top view (not to scale) db7 db6 db5 db4 db3 db2 agnd ldac db0 db1 AD669 ref out ref in v out span/bip offset thd + n ?% temperature ? c 10 0.001 125 0.01 ?5 ?0 0.1 1 100 75 50 25 0 ?0db ?0db 0db thd+n vs. temperature frequency ?hz thd + n ?% 10 0.001 0.01 0.1 1 100 10000 1000 ?0db ?0db 0db thd+n vs. frequency
AD669 rev. a C5C definitions of specifications integral nonlinearity: analog devices defines inte- gral nonlinearity as the maximum deviation of the actual, ad- justed dac output from the ideal analog output (a straight line drawn from 0 to fsC1 lsb) for any bit combination. this is also referred to as relative accuracy. differential nonlinearity: differential nonlinearity is the measure of the change in the analog output, normalized to full scale, associated with a 1 lsb change in the digital input code. monotonic behavior requires that the differential linearity error be within 1 lsb over the temperature range of interest. monotonicity: a dac is monotonic if the output either increases or remains constant for increasing digital inputs with the result that the output will always be a single-valued function of the input. gain error: gain error is a measure of the output error be- tween an ideal dac and the actual device output with all 1s loaded after offset error has been adjusted out. offset error: offset error is a combination of the offset errors of the voltage-mode dac and the output amplifier and is measured with all 0s loaded in the dac. bipolar zero error: when the AD669 is connected for bipolar output and 10 . . . 000 is loaded in the dac, the devia- tion of the analog output from the ideal midscale value of 0 v is called the bipolar zero error. drift: drift is the change in a parameter (such as gain, offset and bipolar zero) over a specified temperature range. the drift temperature coefficient, specified in ppm/ c, is calculated by measuring the parameter at t min , 25 c and t max and dividing the change in the parameter by the corresponding temperature change. total harmonic distortion + noise: total har- monic distortion + noise (thd+n) is defined as the ratio of the square root of the sum of the squares of the values of the har- monics and noise to the value of the fundamental input fre- quency. it is usually expressed in percent (%). thd+n is a measure of the magnitude and distribution of lin- earity error, differential linearity error, quantization error and noise. the distribution of these errors may be different, depend- ing upon the amplitude of the output signal. therefore, to be the most useful, thd+n should be specified for both large and small signal amplitudes. signal-to-noise ratio: the signal-to-noise ratio is de- fined as the ratio of the amplitude of the output when a full- scale signal is present to the output with no signal present. this is measured in db. digital-to-analog glitch impulse: this is the amount of charge injected from the digital inputs to the analog output when the inputs change state. this is measured at half scale when the dac switches around the msb and as many as possible switches change state, i.e., from 011 . . . 111 to 100 . . . 000. digital feedthrough: when the dac is not selected (i.e., cs is held high), high frequency logic activity on the digi- tal inputs is capacitively coupled through the device to show up as noise on the v out pin. this noise is digital feedthrough. theory of operation the AD669 uses an array of bipolar current sources with mos current steering switches to develop a current proportional to the applied digital word, ranging from 0 ma to 2 ma. a seg- mented architecture is used, where the most significant four data bits are thermometer decoded to drive 15 equal current sources. the lesser bits are scaled using a r-2r ladder, then ap- plied together with the segmented sources to the summing node of the output amplifier. the internal span/bipolar offset resistor can be connected to the dac output to provide a 0 v to +10 v span, or it can be connected to the reference input to provide a C10 v to +10 v span. span/ bip off agnd dgnd ref in ref out v out ldac db15 (msb) (lsb) db0 cs l1 ? ee +v cc +v ll 16-bit latch 10k amp AD669 22 7 6 5 23 27 28 10v ref 2 14 3 26 24 10k 10.05k 16-bit latch 16-bit dac 25 figure 2. AD669 functional block diagram analog circuit connections internal scaling resistors provided in the AD669 may be con- nected to produce a unipolar output range of 0 v to +10 v or a bipolar output range of C10 v to +10 v. gain and offset drift are minimized in the AD669 because of the thermal tracking of the scaling resistors with other device components. unipolar configuration the configuration shown in figure 3a will provide a unipolar 0 v to +10 v output range. in this mode, 50 w resistors are tied between the span/bipolar offset terminal (pin 26) and v out (pin 25), and between ref out (pin 28) and ref in (pin 27). it is possible to use the AD669 without any external components by tying pin 28 directly to pin 27 and pin 26 directly to pin 25. eliminating these resistors will increase the gain error by 0.25% of fsr. output gnd ldac db15 (msb) (lsb) db0 cs l1 ? ee +v cc +v ll 16-bit latch 10k amp AD669 22 7 6 5 23 27 28 10v ref 2 1 4 3 10k 10.05k 16-bit latch 16-bit dac 26 25 24 r1 50 w r2 50 w figure 3a. 0 v to +10 v unipolar voltage output
AD669 rev. a C6C if it is desired to adjust the gain and offset errors to zero, this can be accomplished using the circuit shown in figure 3b. the adjustment procedure is as follows: step1 . . . zero adjust turn all bits off and adjust zero trimmer, r4, until the output reads 0.000000 volts (1 lsb = 153 m v). step 2 . . . gain adjust turn all bits on and adjust gain trimmer, r1, until the output is 9.999847 volts. (full scale is adjusted to 1 lsb less than the nominal full scale of 10.000000 volts). output gnd ldac db15 (msb) (lsb) db0 cs l1 16-bit latch 10k v amp AD669 22 7 6 5 23 27 28 10v ref 2 1 4 3 10k v 10.05k v 16-bit latch 16-bit dac 26 25 24 r2 50 v r3 16k v r4 10k v +15v C15v r1 100 v Cv ee +v cc +v ll figure 3b. 0 v to +10 v unipolar voltage output with gain and offset adjustment bipolar configuration the circuit shown in figure 4a will provide a bipolar output voltage from C10.000000 v to +9.999694 v with positive full scale occurring with all bits on. as in the unipolar mode, resis- tors r1 and r2 may be eliminated altogether to provide AD669 bipolar operation without any external components. eliminating these resistors will increase the gain error by 0.50% of fsr in the bipolar mode. r1 50 v output gnd ldac db15 (msb) (lsb) db0 cs l1 16-bit latch 10k v amp AD669 22 7 6 5 23 27 10v ref 2 14 3 26 24 10k v 10.05k v 16-bit latch 16-bit dac 25 28 r2 50 v Cv ee +v cc +v ll figure 4a. 10 v bipolar voltage output gain offset and bipolar zero errors can be adjusted to zero using the circuit shown in figure 4b as follows: step i . . . offset adjust turn off all bits. adjust trimmer r2 to give C10.000000 volts output. step ii . . . gain adjust turn all bits on and adjust r1 to give a reading of +9.999694 volts. step iii . . . bipolar zero adjust (optional) in applications where an accurate zero output is re- quired, set the msb on, all other bits off, and readjust r2 for zero volts output. 100 v r1 output gnd ldac db15 (msb) (lsb) db0 cs l1 16-bit latch 10k v amp AD669 22 7 6 5 23 27 10v ref 2 1 4 3 26 24 10k v 10.05k v 16-bit latch 16-bit dac 25 28 100 v r2 Cv ee +v cc +v ll figure 4b. 10 v bipolar voltage output with gain and offset adjustment it should be noted that using external resistors will introduce a small temperature drift component beyond that inherent in the AD669. the internal resistors are trimmed to ratio-match and temperature-track other resistors on chip, even though their ab- solute tolerances are 20% and absolute temperature coeffi- cients are approximately C50 ppm/ c. in the case that external resistors are used, the temperature coefficient mismatch be- tween internal and external resistors, multiplied by the sensitiv- ity of the circuit to variations in the external resistor value, will be the resultant additional temperature drift. internal/external reference use the AD669 has an internal low noise buried zener diode refer- ence which is trimmed for absolute accuracy and temperature coefficient. this reference is buffered and optimized for use in a high speed dac and will give long-term stability equal or supe- rior to the best discrete zener diode references. the perfor- mance of the AD669 is specified with the internal reference driving the dac since all trimming and testing (especially for gain and bipolar offset) is done in this configuration. the internal reference has sufficient buffering to drive external circuitry in addition to the reference currents required for the dac (typically 1 ma to ref in and 1 ma to bipolar off- set). a minimum of 2 ma is available for driving external loads. the AD669 reference output should be buffered with an external op amp if it is required to supply more than 4 ma total current. the reference is tested and guaranteed to 0.2% max error. the temperature coefficient is comparable to that of the gain tc for a particular grade. if an external reference is used (10.000 v, for example), addi- tional trim range should be provided, since the internal refer- ence has a tolerance of 20 mv, and the AD669 gain and bipolar offset are both trimmed with the internal reference. the optional gain and offset trim resistors in figures 5 and 6 provide enough adjustment range to null these errors. it is also possible to use external references other than 10 volts with slightly degraded linearity specifications. the recom- mended range of reference voltages is +5 v to +10.24 v, which
AD669 rev. a C7C allows 5 v, 8.192 v and 10.24 v ranges to be used. for ex- ample, by using the ad586 5 v reference, outputs of 0 v to +5 v unipolar or 5 v bipolar can be realized. using the ad586 voltage reference makes it possible to operate the AD669 off of 12 v supplies with 10% tolerances. figure 5 shows the AD669 using the ad586 5 v reference in the bipolar configuration. this circuit includes two optional po- tentiometers and one optional resistor that can be used to adjust the gain, offset and bipolar zero errors in a manner similar to that described in the bipolar configuration section. use C5.000000 v and +4.999847 as the output values. 50 w output gnd ldac db15 (msb) (lsb) db0 cs l1 ? ee +v cc +v ll AD669 22 7 6 5 23 27 2 14 3 26 25 r1 100 w r2 10k w 4 6 5 ad586 +v cc 28 ref in ref out v out trim gnd span/bip offset 24 2 figure 5. using the AD669 with the ad586 5 v reference using the AD669 with the ad688 high precision voltage reference the AD669 is specified for gain drift from 15 ppm/ c to 25 ppm/ c (depending upon grade) using its internal 10 volt reference. since the internal reference contributes the vast ma- jority of this drift, an external high precision voltage reference will greatly improve performance over temperature. as shown in figure 6, the +10 volt output from the ad688 is used as the AD669 reference. with a 3 ppm/ c drift over the industrial temperature range, the ad688 will improve the gain drift by a factor of 5 to a factor of 8 (depending upon the grade of the AD669 being used). using this combination may result in ap- parent increases in initial gain error due to the differences between the internal reference by which the device is laser trimmed and the external reference with which the device is ac- tually applied. the AD669 internal reference is specified to be 10 volts 20 mv whereas the ad688 is specified as 10 volts 5 mv. this may result in an additional 5 mv (33 lsbs) of ap- parent initial gain error beyond the specified AD669 gain error. the circuit shown in figure 6 also makes use of the C10 v ad688 output to allow the unipolar offset and gain to be ad- justed to zero in the manner described in the unipolar configuration section. r2 100 w r3 20k r1 100 w ldac db15 (msb) (lsb) db0 cs l1 -v ee +v cc +v ll 16-bit latch 10k amp AD669 22 7 6 5 23 27 28 10v ref 2 1 4 3 26 24 10k 10.05k 16-bit latch 16-bit dac 25 r4 10k w 7 6 3 1 14 15 2 16 5 10 8 12 11 13 r1 r2 r5 r6 r3 ad688 a3 a1 a4 a2 r4 r s 9 4 gnd output 0 to +10v +v cc ? ee figure 6. using the AD669 with the ad688 high precision 10 v reference
AD669 rev. a C8C output settling and glitch the AD669s output buffer amplifier typically settles to within 0.0008% fs (l/2 lsb) of its final value in 8 m s for a full-scale step. figures 7a and 7b show settling for a full-scale and an lsb step, respectively, with a 2 k w , 1000 pf load applied. the guar- anteed maximum settling time at +25 c for a full-scale step is 13 m s with this load. the typical settling time for a 1 lsb step is 2.5 m s. the digital-to-analog glitch impulse is specified as 15 nv-s typi- cal. figure 7c shows the typical glitch impulse characteristic at the code 011 . . . 111 to 100 . . . 000 transition when loading the second rank register from the first rank register. 20 ?0 0 0 +10 10 600 400 200 0 ?00 ?00 ?00 volts m s m v a. C10 v to +10 v full-scale step settling 5 0 0 600 400 200 ?00 ?00 ?00 m s m v 1 234 b. lsb step settling 5 0 0 +10 ?0 m s mv 1 234 c. d-to-a glitch impulse figure 7. output characteristics digital circuit details the bus interface logic of the AD669 consists of two indepen- dently addressable registers in two ranks. the first rank consists of a 16-bit register which is loaded directly from a 16-bit micro- processor bus. once the 16-bit data word has been loaded in the first rank, it can be loaded into the 16-bit register of the second rank. this double-buffered organization avoids the generation of spurious analog output values. the first rank latch is controlled by cs and l1 . both of these inputs are active low and are level-triggered. this means that data present during the time when both cs and l1 are low will enter the latch. when either one of these signals returns high, the data is latched. the second rank latch is controlled by ldac. this input is ac- tive high and is also level-triggered. data that is present when ldac is high will enter the latch, and hence the dac will change state. when this pin returns low, the data is latched in the dac. note that ldac is not gated with cs or any other control sig- nal. this makes it possible to simultaneously update all of the AD669s present in a multi-dac system by tying the ldac pins together. after the first rank register of each dac has been individually loaded and latched, the second rank registers are then brought high together, updating all of the dacs at the same time. to reduce bit skew, it is suggested to leave 100 ns between the first rank load and the second rank load. the first rank latch and second rank latch can be used together in a master-slave or edge-triggered configuration. this mode of operation occurs when ldac and cs are tied together with l1 tied to ground. rising edges on the ldac- cs pair will update the dac with the data presented preceding the edge. the tim- ing diagram for operation in this mode can be seen in figure lb. note, however, that the sum of t low and t high must be long enough to allow the dac output to settle to its new value. table i. AD669 truth table cs l1 ldac operation 0 0 x first rank enable x 1 x first rank latched 1 x x first rank latched x x 1 second rank enabled x x 0 second rank latched 0 0 1 all latches transparent x = dont care it is possible to make the second rank register transparent by ty- ing pin 23 high. any data appearing in the first rank register will then appear at the output of the dac. it should be noted, how- ever, that the deskewing provided by the second rank latch is then defeated, and glitch impulse may increase. if it is desired to make both registers transparent, this can be done by tying pins 5 and 6 low and pin 23 high. table i shows the truth table for the AD669, while the timing diagram is found in figure 1. input coding the AD669 uses positive-true binary input coding. logic 1 is represented by an input voltage greater than 2.0 v, and logic 0 is defined as an input voltage less than 0.8 v.
AD669 rev. a C9C unipolar coding is straight binary, where all zeros (0000h) on the data inputs yields a zero analog output and all ones (ffffh) yields an analog output 1 lsb below full scale. bipolar coding is offset binary, where an input code of 0000h yields a minus full-scale output, an input of ffffh yields an output 1 lsb below positive full scale, and zero occurs for an input code with only the msb on (8000h). the AD669 can be used with twos complement input coding if an inverter is used on the msb (db15). digital input considerations the threshold of the digital input circuitry is set at 1.4 volts. the input lines can thus interface with any type of 5 volt logic. the AD669 data and control inputs will float to indeterminate logic states if left open. it is important that cs and l1 be con- nected to dgnd and chat ldac be tied to v ll if these pins are not used. fanout for the AD669 is 40 when used with a standard low power schottky gate output device. 16-bit microprocessor interface the 16-bit parallel registers of the AD669 allow direct interfac- ing to 16-bit general purpose and dsp microprocessor buses. the following examples illustrate typical AD669 interface configurations. AD669 to adsp-2101 interface the flexible interface of the AD669 minimizes the required glue logic when it is connected in configurations such as the one shown in figure 8. the AD669 is mapped into the adsp- 2101s memory space and requires two wait states using a 12.5 mhz processor clock. in this configuration, the adsp-2101 is set up to use the inter- nal timer to interrupt the processor at the desired sample rate. the wr pin and data lines d8Cd23 from the adsp-2101 are tied directly to the l1 and db0 through db15 pins of the AD669, respectively. the decoded signal cs1 is connected to both cs and ldac. when a timer interrupt is detected, the adsp-2101 automatically vectors to the appropriate service routine with minimal overhead. the interrupt routine then in- structs the processor to execute a data memory write to the ad- dress of the AD669. the wr pin and cs1 both go low causing the first 16-bit latch inside the AD669 to be transparent. the data present in the first rank is then latched by the rising edge of wr . the rising edge of cs1 will cause the second rank 16-bit latch to become transparent updating the output of the dac. the length of w r is extended by two wait states to comply with the timing requirements of t low shown in figure 1b. it is important to latch the data with the rising edge of wr rather than the decoded cs1. this is necessary to comply with the t dh specification of the AD669. a0 d8 adsp-2101 dgnd +5v decoder address bus ldac AD669 dgnd db0 data bus dms wr cs1 cs l1 a13 db15 d23 v ll v ll v out a. adsp-2101 to AD669 interface a13 a12 a11 dms cs1 b. typical address decoder figure 8. adsp-2101 to AD669 interface figure 8b shows the circuitry a typical decoder might include. in this case, a data memory write to any address in the range 3000h to 3400h will result in the AD669 being updated. these decoders will vary greatly depending on the number of devices memory-mapped by the processor. AD669 to dsp56001 interface figure 9 shows the interface between the AD669 and the dsp56001. like the adsp-2101, the AD669 is mapped into the dsp56001s memory space. this application was tested with a processor clock of 20.48 mhz (t cyc = 97.66 ns) although faster rates are possible. an external clock connected to the irqa pin of the dsp56001 interrupts the processor at the desired sample rate. if ac perfor- mance is important, this clock should be synchronous with the dsp56001 processor clock. asynchronous clocks will cause jit- ter on the latch signal due to the uncertainty associated with the acknowledgment of the interrupt. a synchronous clock is easily generated by dividing down the clock from the dsp crystal. if ac performance is not important, it is not necessary for irqa to be synchronous. after the interrupt is acknowledged, the interrupt routine ini- tiates a memory write cycle. all of the AD669 control inputs are
AD669 rev. a C10C tied together which configures the input stage as an edge trig- gered 16-bit register. the rising edge of the decoded signal latches the data and updates the output of the dac. it is neces- sary to insert wait states after the processor initiates the write cycle to comply with the timing requirements t low shown in figure 1b. the number of wait states that are required will vary depending on the processor cycle time. the equation given in figure 9 can be used to determine the number of wait states given the frequency of the processor crystal. a0?15 dsp56001 d0?23 dgnd +5v external clock address decode 74f32 ldac AD669 dgnd db0?b15 xtal cs l1 ds x/ y wr irqa v ll cs1 t = 1 2 (xtal) t ?t + 9ns 2t low # of wait states = v ll figure 9. dsp56001 to AD669 interface as an example, the 20.48 mhz crystal used in this application results in t = 24.4 ns which means that the required number of wait states is about 2.76. this must be rounded to the next highest integer to assure that the minimum pulse widths comply with those required by the AD669. as the speed of the proces- sor is increased, the data hold time relative to cs1 decreases. as processor clocks increase beyond 20.48 mhz, a configuration such as the one shown for the adsp-2101 is the better choice. AD669 to 8086 interface figure 10 shows the 8086 16-bit microprocessor connected to multiple AD669s. the double-buffered capability of the AD669 allows the microprocessor to write to each AD669 individually and then update all the outputs simultaneously. processor speeds of 6, 8, and 10 mhz require no wait states to interface with the AD669. the 8086 software routine begins by writing a data word to the cs1 address. the decoder must latch the address using the ale signal. the decoded cs1 pulse goes low causing the first rank latch of the associated AD669 to become transparent. simultaneously, the 8086 places data on the multiplexed bus which is then latched into the first rank of the AD669 with the rising edge of the wr pulse. care should be taken to prevent excessive delays through the decoder potentially resulting in a violation of the AD669 data hold time (t dh ). the same procedure is repeated until all three AD669s have had their first rank latches loaded with the desired data. a final write command to the ldac address results in a high-going pulse that causes the second rank latches of all the AD669s to become transparent. the falling edge of ldac latches the data from the first rank until the next update. this scheme is easily expanded to include as many AD669s as required. 8086 dgnd +5v ale wr m/ i0 address decode ldac cs1 cs2 cs3 dgnd l1 db0 ?db15 ldac AD669 v out ad0 ?ad15 dgnd l1 db0 ?db15 ldac AD669 v out dgnd l1 db0 ?db15 ldac AD669 cs v out v ll v ll v ll v ll cs cs figure 10. 8086-to-AD669 interface 8-bit microprocessor interface the AD669 can easily be operated with an 8-bit bus by the ad- dition of an octal latch. the 16-bit first rank register is loaded from the 8-bit bus as two bytes. figure 11 shows the configura- tion when using a 74hc573 octal latch. the eight most significant bits are latched into the 74hc573 by setting the latch enable control line low. the eight least sig- nificant bits are then placed onto the bus. now all sixteen bits can be simultaneously loaded into the first rank register of the AD669 by setting cs and l1 low. 8-bit m p and control d7 d0 d7 d0 q7 q0 74hc573 11 ldac msb db8 db7 lsb AD669 cs1 l1 figure 11. connections for 8-bit bus interface
AD669 rev. a C11C noise in high resolution systems, noise is often the limiting factor. a 16-bit dac with a 10 volt span has an lsb size of 153 m v (C96 db). therefore, the noise floor must remain below this level in the frequency range of interest. the AD669s noise spectral density is shown in figures 12 and 13. figure 12 shows the dac output noise voltage spectral density for a 20 v span excluding the reference. this figure shows the l/f corner frequency at 100 hz and the wideband noise to be below 120 nv/ ? hz . figure 13 shows the reference noise voltage spectral density. this figure shows the reference wideband noise to be below 125 nv/ ? hz . 1000 1 11m 100 10 10 100k 10k 1k 100 10m frequency ?hz noise voltage ?nv/ hz figure 12. dac output noise voltage spectral density 1000 1 1 1m 100 10 10 100k 10k 1k 100 10m frequency ?hz noise voltage ?nv/ hz figure 13. reference noise voltage spectral density board layout designing with high resolution data converters requires careful attention to board layout. trace impedance is the first issue. a 306 m a current through a 0.5 w trace will develop a voltage drop of 153 m v, which is 1 lsb at the 16-bit level for a 10 v full-scale span. in addition to ground drops, inductive and ca- pacitive coupling need to be considered, especially when high accuracy analog signals share the same board with digital sig- nals. finally, power supplies need to be decoupled in order to filter out ac noise. analog and digital signals should not share a common path. each signal should have an appropriate analog or digital return routed close to it. using this approach, signal loops enclose a small area, minimizing the inductive coupling of noise. wide pc tracks, large gauge wire, and ground planes are highly recom- mended to provide low impedance signal paths. separate analog and digital ground planes should also be utilized, with a single interconnection point to minimize ground loops. analog signals should be routed as far as possible from digital signals and should cross them at right angles. one feature that the AD669 incorporates to help the user layout is the analog pins (v cc , v ee , ref out, ref in, span/bip offset, v out and agnd) are adjacent to help isolate analog signals from digital signals. supply decoupling the AD669 power supplies should be well filtered, well regu- lated, and free from high frequency noise. switching power sup- plies are not recommended due to their tendency to generate spikes which can induce noise in the analog system. decoupling capacitors should be used in very close layout prox- imity between all power supply pins and ground. a 10 m f tanta- lum capacitor in parallel with a 0.1 m f ceramic capacitor provides adequate decoupling. v cc and v ee should be bypassed to analog ground, while v ll should be decoupled to digital ground. an effort should be made to minimize the trace length between the capacitor leads and the respective converter power supply and common pins. the circuit layout should attempt to locate the AD669, associated analog circuitry and interconnections as far as possible from logic circuitry. a solid analog ground plane around the AD669 will isolate large switching ground currents. for these reasons, the use of wire wrap circuit construction is not recommended; careful printed circuit construction is preferred. grounding the AD669 has two pins, designated analog ground (agnd) and digital ground (dgnd.) the analog ground pin is the high quality ground reference point for the device. any exter- nal loads on the output of the AD669 should be returned to analog ground. if an external reference is used, this should also be returned to the analog ground. if a single AD669 is used with separate analog and digital ground planes, connect the analog ground plane to agnd and the digital ground plane to dgnd keeping lead lengths as short as possible. then connect agnd and dgnd together at the AD669. if multiple AD669s are used or the AD669 shares ana- log supplies with other components, connect the analog and digital returns together once at the power supplies rather than at each chip. this single interconnection of grounds prevents large ground loops and consequently prevents digital currents from flowing through the analog ground.
AD669 rev. a C12C c1555C10C11/91 printed in u.s.a.

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